Frequency modulated signal cancellation in variable power mode for radar applications

ABSTRACT

A radar system operated in a variable power mode includes transmitters, receivers, and a controller. The transmitters transmit digitally modulated signals. The receivers receive radio signals that include transmitted radio signals from the transmitter and reflected from objects in the environment. In addition, an interfering radar signal from a different radar system is received that has been linearly frequency modulated. Each receiver includes a linear frequency modulation canceler that includes a FIR filter, and is configured as a 1-step linear predictor with least mean squares adaptation to attempt to cancel the interfering signal. The prediction is subtracted from the FIR input signal that drives the adaptation and also comprises the canceler output. The controller is configured to control the adaptation on a first receiver. The controller delays the adaptation such that transients at the start of each receive pulse are avoided.

CROSS REFERENCE TO RELATED APPLICATIONS

The present application claims the filing benefits of U.S. provisionalapplication, Ser. No. 62/598,563, filed Dec. 14, 2017, which is herebyincorporated by reference herein in its entirety.

FIELD OF THE INVENTION

The present invention is directed to radar systems, and moreparticularly to radar systems for vehicles.

BACKGROUND

The use of radar to determine range, angle, and velocity of objects inan environment is important in a number of applications includingautomotive radar and gesture detection. A radar system typicallytransmits a signal and “listens” for the reflection of the signal fromobjects in the environment. By comparing the transmitted signal with thetiming of the received signal, a radar system can determine the distanceto an object. By observing the Doppler shift in the frequency of thereflected signal relative to the transmitted signal, the velocity of theobject can also be determined. Moreover, by using multiple transmittersand/or receivers, the location (angle) of the object can also bedetermined.

There are several types of waveforms used in different types of radarsystems. One type of waveform or radar signal is known as afrequency-modulated continuous waveform (FMCW). In an FMCW-type radarsystem, the transmitter of the radar system sends a continuous signal inwhich the frequency of the signal varies. This is sometimes called achirp radar system. Mixing (multiplying) a waveform reflected from anobject (also known as a target) with a replica of the transmitted signalresults in a CW signal with a frequency that represents the distancebetween the radar transmitter/receiver and the target. By sweeping up infrequency and then down in frequency, the Doppler frequency can also bedetermined. Another type of waveform used in a radar system is known asa phase-modulated continuous waveform (PMCW). In a PMCW-type radarsystem the transmitter of the radar system sends a continuous signal inwhich the phase of the signal varies. By filtering the received signalwith a filter matched to the transmitted signal the autocorrelation ofthe signal is generated. This will have large magnitude peaks at timedelays corresponding to the round trip distance between the transmitterand receiver.

SUMMARY

Methods and systems of the present invention include a PMCW radar systemthat includes transmitters for transmitting radio signals that are phasemodulated, and receivers for receiving radio signals that includetransmitted radio signals transmitted by the transmitters and reflectedfrom objects in the environment. The received radio signals includefrequency modulated signal interference from other sources. Theirfrequency modulated signals deviate the carrier signal frequency from amean or center frequency according to linear frequency transitions. Goodperformance in mitigating frequency modulated signal interference may beachieved with reduced transient levels at the beginning of each receivepulse by delaying adaptation of a linear frequency modulation canceler(FCU), such that coefficients of finite impulse response (FIR) filtersof the FCU are filled with valid signal samples before the adaptation.During the delay, a training pass may be used to supply the valid signalsamples. Each of the receivers will include an FCU for mitigating thefrequency modulated signal interference.

In accordance with an aspect of the present invention, a radar systemfor a vehicle includes transmitters, receivers, and a controller forcontrolling the transmitters and receivers. The transmitters transmitphase-modulated radio signals. The receivers are configured to receiveradio signals that include transmitted radio signals transmitted by thetransmitters and reflected from objects in the environment. Thereceivers are further configured to receive frequency modulated signalinterference transmitted by another source. Each receiver of theplurality of receivers includes a linear frequency modulation cancelerconfigured to cancel or mitigate any frequency-modulated signalinterference. A linear frequency modulation canceller includes a FIRfilter and is configured as a 1-step linear predictor with least meansquares adaptation. The prediction is subtracted from the FIR inputsignal that drives the adaptation and also comprises the canceleroutput.

In accordance with an aspect of the present invention, a method formitigating frequency modulated interference in a radar sensing systemincludes providing a plurality of transmitters configured forinstallation and use on a vehicle. A plurality of receivers are providedthat are configured for installation and use on the vehicle.Phase-modulated radio signals are transmitted with the transmitters.Radio signals are received with the receivers. The received radiosignals include at least one of: transmitted radio signals transmittedby the transmitters and reflected from objects in the environment, andinterfering frequency-modulated radio signals transmitted by anotherradar sensing system. The transmitters and receivers are controlled suchthat the transmitters and the receivers operate in alternating transmitand receive windows. Each receiver of the plurality of receiverscomprises a linear frequency modulation canceler. A linear frequencymodulation canceler mitigates any interfering frequency-modulatedsignals received by an associated receiver, such that the associatedreceiver is able to estimate a location of an object.

In accordance with an aspect of the present invention, a radar sensingsystem includes a transmitter, a receiver, and a controller. Thetransmitter is configured to transmit phase-modulated continuous waveradio signals. The receiver is configured to receive radio signals thatinclude (i) the transmitted radio signals transmitted by the transmitterand reflected from objects in the environment, and (ii) interferingfrequency-modulated radio signals transmitted by another radar sensingsystem. The controller is configured to control the transmitter and thereceiver such that the transmitter and the receiver operate inalternating transmit and receive windows. The receiver also includes afrequency modulation canceller configured to cancel a substantialportion of the interfering radio signals received by the receiver, suchthat the receiver is able to estimate a location of an object.

In an aspect of the present invention, the controller is configured tocontrol the adaptation on a first receiver. Optionally, FIR coefficientsfor all the other receivers of the plurality of receivers are copiedfrom the first receiver. The controller may also delay the adaptationsuch that transients at the start of each receive pulse are avoided.

In accordance with a further aspect of the present invention, thecontroller is configured to delay the FIR adaptation until a delay lineof the FIR filter has filled with valid signal samples. Before the FIRadaptation begins, a training pass is performed to fill FIR filtercoefficients with the samples. Furthermore, the training pass is runbackwards so that the FIR filter coefficients at the end of the trainingcorrespond to a chirp frequency at the beginning of the pulse.

These and other objects, advantages, purposes and features of thepresent invention will become apparent upon review of the followingspecification in conjunction with the drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a plan view of an automobile equipped with one or more radarsystems in accordance with the present invention;

FIG. 2A and FIG. 2B are block diagrams of radar systems in accordancewith the present invention;

FIG. 3 is a block diagram illustrating a flow of data structures througha radar system in accordance with the present invention;

FIG. 4 is a block diagram illustrating a radar system with a pluralityof receivers and a plurality of transmitters (MIMO radar) for producingthe data structures of FIG. 3, in accordance with the present invention;

FIG. 5 is a graph illustrating an output of a simulated finite impulseresponse (FIR) adaptation in a continuous power mode in accordance withthe present invention;

FIG. 6 is a graph illustrating exemplary setting curves with a constantfrequency CW tone in accordance with the present invention;

FIG. 7 is a graph illustrating a settling transient caused by starting aFIR adaptation before a delay line of the FIR filter has filled withvalid signal samples in accordance with the present invention;

FIGS. 8 and 9 are graphs illustrating the settling of FIR coefficientswithout and with a hold-off, respectively, to freeze adaptation until aFIR delay line is filled in accordance with the present invention;

FIGS. 10 and 11 are graphs illustrating exemplary start-up transientswith extended settling times in accordance with the present invention;

FIG. 12 is a graph illustrating a long settling time of FIR coefficientsthat is obscured by coefficient tracking of a frequency modulated chirpin accordance with the present invention;

FIG. 13 is a graph illustrating an exemplary adaptive canceller learningcurve in accordance with the present invention;

FIG. 14 is a graph illustrating a variable power mode modeled where asettling transient appears at the beginning of each receive window inaccordance with the present invention;

FIG. 15 is a graph illustrating an effect of enabling an adaptationhold-off during delay line filling in accordance with the presentinvention;

FIG. 16 is a graph illustrating an exemplary error learning curve with abackwards training section prior to each pulse in accordance with thepresent invention;

FIG. 17 is a graph illustrating a mu of 2⁻⁹ and an 80-sample look-aheadin accordance with the present invention;

FIG. 18 is a graph illustrating a mu of 2⁻¹⁰ and a 200-sample look-aheadin accordance with the present invention;

FIG. 19 is a graph illustrating the settling of coefficients for arepresentative pulse during a backward training pass in accordance withthe present invention;

FIG. 20 is a graph illustrating additional amplitude plots in accordancewith the present invention;

FIG. 21 is a graph illustrating the plotting of FIR zeroes at the end ofa simulation in accordance with the present invention;

FIG. 22 is a graph of exemplary frequency plots illustrating theposition of a canceller notch at the end of an FMCW ramp in accordancewith the present invention;

FIG. 23 is a graph illustrating coefficient tracking in accordance withthe present invention;

FIG. 24 is a block diagram illustrating basic processing blocks of atransmitter and receiver in an exemplary radar system in accordance withthe present invention;

FIG. 25 is a block diagram illustrating an exemplary baseband signalprocessor in accordance with the present invention; and

FIG. 26 is a block diagram illustrating an adaptive filter forfrequency-modulated signal interference mitigation in accordance withthe present invention.

DETAILED DESCRIPTION

The present invention will now be described with reference to theaccompanying figures, wherein numbered elements in the following writtendescription correspond to like-numbered elements in the figures. Methodsand systems of the present invention may achieve a good performance inmitigating frequency-modulated signal interference with reducedtransient levels at the beginning of each receive pulse or window bydelaying adaptation of a linear frequency modulation canceler (FCU),such that coefficients of FIR filters of the FCU are filled with validsignal samples before the adaptation.

There are several different types of radar systems. The most well-knownis pulse radar, in which a very short pulse of very high-power microwaveenergy is transmitted during which time the receiver is blanked toprevent overload or damage; then the receiver is unblanked and listensfor echoes received with various delays. The length of time the receivercan listen before the next transmitter pulse equates to the maximumrange. The antenna may rotate between pulses to test for reflectingobjects at different azimuths or elevations or both.

A less common variation of the above is the bistatic radar system inwhich the transmitter is not co-located with the receiver and uses atotally different antenna. The receiver thereby does not need to beblanked during the transmit pulse.

In pulse radar systems, the transmitter duty factor and therefore themean power is small; therefore, to achieve enough sensitivity for longrange performance a high peak pulse power must be used. To overcomethat, another type of radar called continuous wave (CW) radar is used. ACW radar transmits and receives all the time. The transmitted signal hasfeatures in its waveform that enable the receiver to determine the delayof a received signal by determining the time difference between thetransmitted feature and the received feature. In FMCW-type radarsystems, the feature used is the instantaneous frequency. Thetransmitter frequency is changed linearly and very rapidly from astarting value to an ending value to create what is known as a chirp. Adelayed signal will be received at an earlier value of the chirpfrequency. By forming a beat between the transmit frequency and thereceived frequency in the receive mixer, and determining the beatfrequency, which is the transmit-receive frequency difference, the delayof the reflected chirp can be calculated. Because such a frequencydifference cannot be distinguished from Doppler, a forward and backwardchirp may be used alternately, producing a sawtooth frequencymodulation. Any Doppler has an opposite effect on interpreting theforward chirp compared to the backward chirp, thus allowing range andDoppler to be separated. In FMCW radar systems, one issue is the extremeaccuracy and linearity needed for the chirp signal. The greatest issuein CW radar is receiving at the same time as transmitting. Thetransmitted signal is much stronger than any received echo and canoverload the receiver's limited dynamic range. Improving the frequencymodulation performance of FMCW-type radar systems is described in detailin U.S. Publication No. US-2017-0307728, which is hereby incorporated byreference herein in its entirety.

Another version of CW radar called pulse-CW radar aims to reduce thedifficulty of receiving weak echoes from distant objects in the presenceof the strong own transmitter signal. This is similar to pulse radarexcept that the transmitter duty factor is much higher, for example 50%.A modulated transmit pulse is transmitted for a duration that fills upthe time to the furthest object and then switches off. The receiverreceives strong echoes from nearby objects while the transmitter istransmitting, but when weak echoes from distant objects are received,the transmitter has already switched off so that there are no signalsbeing received from nearby objects. This facilitates detection ofdistant objects. Improving near-far performance in radar systems isdescribed in detail in U.S. Pat. No. 9,753,121 (“the '121 patent”),which is hereby incorporated by reference herein in its entirety.

The invention is described primarily for use in a digital PMCW radar inwhich transmission and reception occur alternately at a same site andthe interference is an FMCW type of radar. As discussed herein,FMCW-type radars may also be operated such that transmission andreception occur simultaneously. Hybrid radars can also be made in whichtransmission and reception are simultaneous for a first period and thenthe transmitter switches off to allow the receiver to receive weak, lateechoes without strong interference from the local transmitter, asdiscussed in the '121 patent. As noted above, FMCW radar typically usedchirp signals to determine range and Doppler.

Radars with a single transmitter and a single receiver can determinedistance to a target but cannot accurately determine a direction or anangle of a target from the radar sensor or system unless the antennapattern is steered between pulses either mechanically or electronicallyusing a phased-array. To acquire angular information for each radarpulse period, which in the case of the exemplary radar system describedherein comprises a sequence of frequency modulating bits with which thereceiver performs correlation, either multiple transmitter antennas ormultiple receiver antennas or both are needed, and which are operativein all directions all the time. Each receiver receives and separateseach echoed transmitter signal, thus resulting in N x M receivedresults, where N is the number of transmitters and M is the number ofreceivers. These signals can be individually processed to determinerange and velocity. With proper design, these N x M signals fromdifferent virtual receivers/radars can be combined in any number of waysaccording to a plurality of beamforming vectors, thereby achievingelevation and azimuth location of each object as well as range andDoppler information.

The larger the number of transmitter antennas and receiver antennas, thebetter the resolution possible. Each transmission antenna is connectedto a separate transmitter, and each receiver antenna is connected to aseparate receiver. As discussed herein, such a radar system is known asa multiple-input, multiple-output (MIMO) radar system.

An exemplary MIMO radar system is illustrated in FIG. 4. With MIMO radarsystems, each transmitter signal is rendered distinguishable from everyother transmitter by using appropriate differences in the modulation,for example, different digital code sequences. Each receiver correlateswith each transmitter signal, producing a number of correlated outputsequal to the product of the number of receivers with the number oftransmitters. The outputs are deemed to have been produced by a numberof virtual receivers, which can exceed the number of physical receivers.A receiver may be referred to as a virtual receiver even when there isonly a single transmitter, in order to avoid changing the terminology.The output of a given receiver receiving a transmitted signal from agiven transmitter has a phase that depends on the round-trip distancefrom the transmitting antenna to the receiving antenna. Eachtransmit-receive combination produces a different phase due to theseparation of their antennas. By combining the outputs for eachtransmitter/receiver combination while correcting for these differentphase shifts, a combined output is obtained that only constructivelyadds for a target at a unique point in space. By repeating thecombination using the precomputed phase shifts for many different pointsin space, signals may be resolved in the three dimensions of range,azimuth, and elevation. The focusing effect of the above phase coherentcombining is effective for resolution in azimuth and elevation but onlycontributes to range resolution at very short ranges, and the rangeresolution at long ranges is principally determined by the round-tripdelay of the digital modulation. An exemplary radar system according tothe invention therefore determines the range of a target or the distanceto a target principally by determining how long it takes an echo oftransmitted RF signals to be heard back at the receivers. From thismeasured time-delay and knowing that the electromagnetic RF signalstravel at the speed of light (or ultrasonic signals traveling at thespeed of sound), the distance can be determined.

In digital PMCW radar, the method of determining the time delay isaccomplished by correlating a received RF signal with multipletime-shifts of the digital modulating code to produce correlations whichare stored in range bins. The length of time over which coherentcorrelations can be performed is limited by the phase rotation caused byDoppler shift. To continue cumulative correlation for longer times thanthis, partial correlations are combined while compensating for theDoppler-induced phase drift. The partial correlations may be stored foreach virtual receiver and range in a 3-dimensional array called a radardata cube, as illustrated in FIG. 3, in which the three dimensions arevirtual receiver number, range, and time or index of the partialcorrelation. Partial correlations for the same receiver and range arethen submitted to an FFT, which combines them in a computationallyefficient manner with many different hypotheses of the rate-of-change ofphase, thus producing long correlations for each of a number of Dopplerbins. The result is then stored in a radar data cube having thedimensions of virtual receiver number, range, and Doppler shift. Thus,the radar data cube time dimension has been converted into a Dopplershift dimension which is more meaningful for characterizing a reflectingtarget or object as stationary or moving. Then, for the same range andDoppler bin, the results across different virtual receivers may becombined by using beamforming matrices as mentioned above in order toachieve angular resolution in azimuth, elevation or both.

Because there can be multiple objects in the environment, there will bemultiple bins in the radar cube for which there will be a highcorrelation. While a virtual receiver/radar could correlate the receivedRF signal with all possible delays, generally there is a finite set ofdelays with which the virtual receiver/radar will correlate, that is, afinite set of range bins over the range of interest. Likewise, therewill be a finite set of Doppler bins up to the maximum conceivablerelative velocity between the radar and an oncoming vehicle. Because thetransmission and return range changes at twice the relative velocity ofthe target to the radar, the maximum Doppler shift may be based oneither twice the maximum speed of any one vehicle if the vehicle isapproaching the radar or negative twice the maximum speed if the vehicleis going away from the radar. The possible range of Doppler is then fourtimes the maximum speed of a vehicle. For a maximum vehicle speed of 250km/hr, which can be reached on the German Autobahn for example, themaximum range of Doppler shift can be 1000 km/hr, which is 74 KHz at 80GHz. If a radar system's own velocity, which is presumed to be known, isdigitally removed by applying a systematic phase de-twisting to thereceived data, the maximum range of Doppler shift drops to 37 KHz.

The radar sensing system of the present invention may utilize aspects ofthe radar systems described in U.S. Pat. Nos. 9,846,228; 9,806,914;9,791,564; 9,791,551; 9,772,397; 9,753,121; 9,599,702; 9,575,160 and/or9,689,967, and/or U.S. Publication Nos. US-2017-0309997; US-2017-0307728and/or US-2017-0310758, and/or U.S. patent applications, Ser. No.15/496,038, filed Apr. 25, 2017, and/or Ser. No. 15/705,627, filed Sep.15, 2017, and/or U.S. provisional application Ser. No. 62/528,789, filedJul. 5, 2017, which are all hereby incorporated by reference herein intheir entireties.

FIG. 1 illustrates an exemplary radar system 100 configured for use in avehicle 150. A vehicle 150 may be an automobile, truck, or bus, etc. Asillustrated in FIG. 1, the radar system 100 may comprise one or moretransmitters and one or more receivers 104 a-104 d which can be usedjointly to realize a plurality of virtual receivers/radars. Otherconfigurations are also possible. FIG. 1 illustrates a radar system 100comprising one or more receivers/transmitters 104 a-104 d, control andprocessing module 102 and indicator 106. The receivers/transmitters 104a-104 d are placed to acquire and provide data for object detection andadaptive cruise control. The radar system 100 (providing such objectdetection and adaptive cruise control or the like) may be part of anAdvanced Driver Assistance System (ADAS) for the automobile 150.

FIG. 2A illustrates an exemplary radar system 200 with an antenna 202that is time-shared between a transmitter 206 and a receiver 208 via aduplexer 204. As also illustrated in FIG. 2A, output from the receiver208 is received by a control and processing module 210 that processesthe output from the receiver 208 to produce display data for the display212. The control and processing module 210 is also operable to produce aradar data output that is provided to other control units. The controland processing module 210 is also operable to control the transmitter206.

FIG. 2B illustrates an alternative exemplary radar system 250 with apair of antennas 202 a, 202 b: an antenna 202 a for the transmitter 206and another antenna 202 b for the receiver 208. While pulse radarsystems may use shared or separate antennas, continuous-wave radars(discussed herein) will use separate antennas (for transmitting andreceiving) because of their continuous operation. Despite usingdifferent antennas, local spillover from transmitter to receiver is ahuge signal having a short delay. A critical issue in CW radar is theremoval by subtraction of this large local spillover signal, for thesuccess of which an accurately defined modulation, as is disclosedherein, is essential.

FIG. 4 illustrates an exemplary digitally-modulated continuous-waveradar system 400. Radar system 400 comprises a plurality of receiversand their respective antennas 406 and a plurality of transmitters andtheir respective antennas 408. The radar system 400 also includes aflash memory 412, and optionally a random-access memory 410. The randomaccess memory 410, for example, an external DRAM, may be used to storeradar data cube(s) instead of using the limited internal (on-chip)memory (e.g., SRAM), and may also be used to store selected range binsfrom a greater number of radar data cubes for post processing to improveDoppler resolution or range resolution by tracking objects using Kalmanfiltering. The radar system may also include a variety ofinterconnections to an automotive network, e.g., Ethernet, CAN-FD,and/or Flexray.

Often it is necessary to transmit a signal for a certain time durationand then turn off or reduce the power of the transmitter. This is sosignals reflected from far objects that are very small are not dominatedby large signals caused by reflections from objects closer to the radarthan the far objects. This mode of operation, known as variable powermode, can help in the detection of the far objects in the presence ofnear objects. The variable power mode includes turning the receiver offwhile the reflections from the near objects are being received and thenturning the transmitter down or off for a time duration so that thesignals from the far objects can be received.

In addition to being able to detect far objects in the presence of nearobjects, it is also possible that a radar system can be subject tointerference from other radars, either of the same type or differenttypes. For example, an interfering radar of the FMCW type might beoperating and causing interference to a radar system of the PMCW type.While techniques exist for mitigating the effect of an FMCW interferingradar on a PMCW victim radar where the receiver is operatingcontinuously, there is a continuous need for improved radar techniquesthat achieve good interference mitigation performance when operating aPMCW-type radar system, such as in a variable power mode.

One significant problem with automotive radar systems is interferencefrom other radar systems in other vehicles. Other radar systems maytransmit an FMCW type of signal and cause interference to a victimradar. Because the radar system on another vehicle has only a one-waypropagation loss, compared to a two-way or round-trip propagation lossof the signal from the victim radar to an object and then back, thereceived signal from an interfering radar might be much larger in powerthan the signal the victim radar is trying to use to estimate locationof objects in the environment. A method of reducing the interference ofan FMCW interfering radar was disclosed in U.S. Pat. No. 9,791,564 B1(“the '564 Patent”), which is incorporate herein in the entirety.

An exemplary block diagram of a radar configured for mitigating an FMCWinterfering signal is illustrated in FIG. 24. A baseband signalgenerator 510 is configured to generate a digital signal which isconverted to an analog signal by a digital-to-analog converter 520 andthen upconverted (530) before being transmitted by an antenna 540. Atthe receiver 550 the signal from the antenna 560 is down-converted andthen converted into a digital signal by an ADC 580 before beingprocessed digitally by a processor 590. The digital signal processing,illustrated in FIG. 25, includes an FMCW cancellation unit (FCU) 610followed by correlation and FFT processing (620). The FCU unit 610 isillustrated in FIG. 26, in which a finite impulse response (FIR) filter810 processes the samples to attempt to notch out the signal of the FMCWinterferer. An error signal is generated at the output of an adder 820and is used to update the weights (830) used in the filter. This loop(810, 820, 830, and 840) implements the adaptive filtering. The filter810 is adaptively updated because the frequency of the FMCW signal ischanging with time.

An exemplary radar sensing system with a linear frequency modulation(LFM) canceler (FCU) may consist of a complex-tap finite impulseresponse (FIR) filter per receiver path, configured as a 1-step linearpredictor with least mean squares (LMS) adaptation. The programmabilityis limited to a maximum number of taps, with the preferred embodimentbeing a maximum of 15 taps. The prediction is then subtracted from theFIR input signal, which forms the error signal that drives the adaptionand also comprises the FMCW canceller output (thus, effectively a 21-tapFIR with one coefficient fixed at 1.0). The adaption occurs only on onereceiver path, and the FIR coefficients for all other receiver paths arecopied from the adaptive path.

The '564 patent is directed towards a continuous operation of a receiverwhereby the receiver is continuously processing the received signal.Because of the near-far problem whereby echoes received from nearbytargets can drown out echoes received from distant targets (andinterfere with the detection of far targets), an exemplary variablepower mode can be employed whereby the transmitter power level is notconstant. A particular realization of a variable power mode is where thetransmitter is either ON or OFF. When the transmitter is OFF, and afterthe echoes from nearby objects/targets have been received, echoes fromdistant objects/targets can be received/detected without the echoes fromthe nearby objects interfering. Exemplary variable power modes can alsoinclude separate operational windows for both transmitter(s) andreceiver(s). However, solutions developed for cancelling or mitigatingfrequency-modulated signal interference when the victim radar system isunder continuous operation do not work as well when the victim radarsystem is operated in a variable power mode. The problem is that thefrequency of an FMCW chirp that is to be cancelled will continue tochange during any time when the receiver is disabled (and the receiveris blind), such that when the receiver starts operating again, thisfrequency error will cause a transient during which error residuals arelarge (cancellation is ineffective) until the adaptation re-settles.

Before addressing variable power modes, system behavior in a continuouspower mode is discussed. The output of a simulated FIR adaptation incontinuous power mode is illustrated in FIG. 5. The signal is wellsettled, in terms of error residual, within 30 samples. The residualsafter that (which actually increase slightly from time index 30 throughabout time index 500 to a final size of about −40 dB) are due to theimperfect ability of the adaptation to follow the frequency change ofthe interfering signal as the FM chirp progresses. There is somefrequency tracking lag, where the FIR notch is always a little bitbehind the actual frequency of the interfering FMCW chirp as it ramps.FIG. 6 illustrates the settling curves with a constant frequency CWtone, which eliminates the effect of the lag on the notch tracking thechirp.

So, without tracking lag of the FM chirp, the final error floor is about−52 dB, which is lower than with the chirp enabled, and is achieved inabout 30 samples of settling time. The level of the error floor (in thiscase, the −52 dB) is governed by a balance between the leakagecoefficient (here set to 0.999) and the loop gain, mu (here set to0.02). One contribution to the settling transient in this simulation iscaused by starting the FIR adaptation before the 20-tap delay line ofthe FIR filter has filled with valid signal samples. This is illustratedin FIG. 7, where the variable hold_adapt_for_pipe_fill is set to TRUE inan updated MATLAB script.

More telling is to look at the settling of the FIR coefficientsthemselves, illustrated in FIGS. 8 and 9, where the settling isillustrated without and then with a hold-off that freezes adaptationuntil the FIR delay line is filled, respectively.

As can be seen by comparing the two plots of FIGS. 8 and 9, allowingadaptation while the FIR delay line is filling causes a start-uptransient in the FIR coefficient values which takes a very long time(several thousand samples) to settle out, but is pretty much invisiblein the error residual curve. This can be attributed to the fact that oneof the zeros (roots) of the FIR filter settles very quickly to notch outthe CW interfering tone, while all the rest of the FIR zeros (roots)take some time to settle to their final positions. There is not reallymuch error energy to drive these zeros to any particular position. Thisis due possibly to numerical noise on the order of digital quantizationerrors. The same basic behavior with the same basic settling time can beobserved even when some white Gaussian noise (WGN) is added to thesimulated. This is illustrated in FIGS. 10 and 11.

When the interference is a chirp instead of a constant-frequency tone,the long settling time of the coefficients due to the pipe-fillingtransient is mostly obscured by the coefficient tracking of the FMchirp, illustrated in FIG. 12.

Loop Delay:

In an exemplary simulation, the effect of register-transfer level (RTL)pipeline stages are not included in the exemplary adaptation loop (810,820, 830, and 840 of FIG. 26). This means that all of the continuouspower mode simulation plots (FIGS. 5-12) are idealized. The amount ofRTL pipelining delay in an exemplary RTL is 13 clock cycles. In a worstcase, the clock rate and symbol rate will be the same (and the FCUnormally operates at symbol rate) so the loop delay is 13 samples. Forlower symbol rates, the loop delay may be fewer samples. When arealistic loop delay is added to the model (e.g., a model created usingmathematical modeling software, such as MATLAB®), it becomes necessaryto turn down the adaptive loop gain (mu) in order to preserve stability.A settling of mu=0.002 (reduced by a factor of 10) seems to providealmost as much stability margin (but not quite, its slightlyunderdamped) as 0.02 did without the loop delay. The leakage factor of0.999 seems to still be suitable. FIG. 13 illustrates an exemplaryadaptive canceller learning curve for this case.

Note that the settling time has increased from 30 samples to 70 (or 120,depending on what is considered “settled”). Also note that thecancellation never gets better than about −20 dB, compared to −40 dBwithout the loop delay in the model. The latter is primarily because thesmaller mu has more trouble tracking the fast frequency change of thischirp.

Hereinafter, a loop delay of 13 samples will be included. In oneexemplary embodiment, a variable power mode is modeled (e.g., in aMATLAB script) by zeroing out 256 out of every 512 signal samples andfreezing adaptation during this time. This causes a settling transientto appear at the beginning of each receive window, as illustrated inFIG. 14.

FIG. 15 illustrates the effect of enabling the adaptation hold-offduring delay line filling. As illustrated in FIG. 15, enabling theadaptation hold-off during delay line filling changes the picture alittle, but does get rid of the transients at the start of each receivepulse.

A proposed solution is to make an initial training pass with the datafrom each pulse before producing output to the downstream blocks.Because the frequency of the LFM chirp changes over the pulse, thetraining pass is run backwards so that the state of the FIR filtercoefficients at the end of the training corresponds to the chirpfrequency at the beginning of the pulse. It turns out that only aportion of the pulse needs to be used in the backwards training pass.The error learning curve with the backwards training section prior toeach pulse is illustrated in FIG. 16.

As illustrated in FIG. 16, the transients are now much smaller and nolonger dominate the MSE averaged over the pulse. Part of the secret tothe absence of big transients in the above plot is to make sure thedelay line of the FIR filter starts with a “valid” signal when forwardoperation begins after the backward training pass. For the first FIRoutput sample, the delay line should be filled with received samplesfrom prior to the beginning of the receive window (i.e., that was neveractually received). These samples can be estimated, however, by runningthe backward training pass beyond the beginning of the receive pulse(into negative look-ahead time indices) by an amount equal to the lengthof the FIR delay line (20 samples).

The training time (number of samples required in the backward trainingpass) depends on the adaptation loop gain, mu. And in turn, theappropriate setting for mu involves some tradeoffs. A larger mu helpswith tracking a fast-changing chirp frequency with less lag and greatersuppression, and helps reduce the settling time (and backwards trainingtime). A smaller mu helps to track a noisy tone more accurately, insituations where the tone is not too far above the noise floor. At verylarge values of mu, the adaptive loop becomes unstable. All thingsconsidered, it appears that the most interesting mu settings with 13samples of loop delay are 2⁻⁹ or 2⁻¹⁰. At 2⁻¹⁰, 200 look-ahead samplesare required in the backward training pass to achieve the performancepotential. At 2⁻⁹, 80 look-ahead samples are required. In either case,the number of operation cycles of the backward predictor is 20 (lengthof the FIR) more than the quoted look-ahead because of the part at theend of the training pass where the “artificial history” is constructedto initialize the FIR delay line. When the symbol rate is lower than theclock rate, and hence the loop delay as measured in FCU samples is lowerthan 13, some larger values of mu may be useable (up to about 2⁻⁵ withzero delay) and a smaller look-ahead may be adequate. This simulationwith mu=2⁻⁹ and an 80-sample look-ahead is illustrated in FIG. 17, withan SNR of 30 dB on the FMCW tone.

Note that the interfering FMCW signal has been attenuated byapproximately 20 dB. Changing mu to 2⁻¹⁰ and with a 200-samplelook-ahead, the result is illustrated in FIG. 18.

Therefore, the reduction in mu has resulted in about 4 dB lessattenuation of the FMCW interferer. However, the smaller mu may performbetter when the FMCW chirp frequency ramp is slower, and/or the SNR islower.

The settling of the coefficients for a representative pulse during thebackward training pass corresponding to the same test conditions asillustrated in FIG. 18, is illustrated in FIG. 19.

The other plots produced by the model (e.g., MATLAB scripts) for thesame simulation as the two plots discussed above, are illustrated inFIG. 20 for completeness.

Note that the FIR zeros plotted in FIG. 21 represent the end of thesimulation, after the completion of the chirp ramp in frequency. Theright-hand plot is simply the FFT of a windowed portion of the inputsignal near the end of the simulation run, so it shows the interferingtone itself, with some spectral spreading due to the observationwindowing inherent in variable power mode. The frequency response plotsillustrated in FIG. 22 are also taken from the end of the simulationrun, so they illustrate the position of the canceller notch at (actuallynear, not at) the end of the FMCW ramp.

The coefficient tracking illustrated in FIG. 23 has counter-intuitivestep functions in coefficient value from one receive pulse window to thenext. This is due to the fact, discussed herein, that the “other” zerosof the FIR filter settle much more slowly than the “important” zero thatnotches the FMCW tone, and consequently the FIR coefficients continue todrift as these other zeros move around (in addition, it appears thatthese “unused” zeros may want to settle to different locations duringthe backward pass as compared to the forward operation). Althoughpsychologically unsettling to an astute observer, this effect does notseem to have a detrimental impact on the cancellation performance of thearchitecture.

The exemplary embodiments disclosed herein cover many variations of PMCWradar systems. As discussed herein, an exemplary PMCW radar system withfrequency modulated interference mitigation includes a linear frequencymodulation canceller (FCU) that includes a FIR filter per receive path,and is configured as a linear predictor with an LMS adaptation. Theprediction is subtracted from the FIR input signal, which forms theerror signal that drives the adaption and also comprises an FMCWcanceler output. Reduced transient levels at the beginning of eachreceive pulse may be achieved by delaying the adaptation, such thatcoefficients of FIR filters are filled with valid signal samples beforethe adaptation, and a person of normal skill in the art can derive manyother variations using the principles exposed herein without departingfrom the spirit and scope of the invention as described by the attachedclaims.

Changes and modifications in the specifically described embodiments canbe carried out without departing from the principles of the invention,which is intended to be limited only by the scope of the appendedclaims, as interpreted according to the principles of patent lawincluding the doctrine of equivalents.

The invention claimed is:
 1. A radar sensing system comprising: at leastone transmitter configured for installation and use on a vehicle, andconfigured to transmit phase-modulated continuous wave radio signals; atleast one receiver configured for installation and use on the vehicle,and configured to receive radio signals that include (i) the transmittedradio signals transmitted by the at least one transmitter and reflectedfrom objects in the environment, and (ii) interferingfrequency-modulated radio signals transmitted by another radar sensingsystem; a controller configured to control the at least one transmitterand the at least one receiver such that the at least one transmitter andthe at least one receiver operate in alternating transmit and receivewindows, wherein the transmit window and the receive window areseparated in time; wherein each receiver of the at least one receivercomprises a linear frequency modulation canceler configured to mitigatethe interfering radio signals received by the associated receiver, suchthat the associated receiver is able to estimate a location of anobject; and wherein the linear frequency modulation canceler comprises afinite impulse response (FIR) filter and is configured as a 1-steplinear predictor with least mean squares adaptation, wherein theprediction is subtracted from an FIR input signal to generate an errorsignal that drives the adaptation, and wherein the linear frequencymodulation canceler generates the canceler output.
 2. The radar sensingsystem of claim 1, wherein the controller is configured to control theadaptation on a first receiver, and wherein FIR coefficients for anyother receivers of the at least one receiver are copied from the firstreceiver, wherein the adaptation is a FIR adaptation.
 3. The radarsensing system of claim 1, wherein the controller delays the adaptationsuch that transients at the start of each receive window are avoided. 4.The radar sensing system of claim 1, wherein the FIR filter comprises amaximum of 15 taps.
 5. The radar sensing system of claim 3, wherein thecontroller is configured to delay the FIR adaptation until a delay lineof the FIR filter has filled with valid signal samples, and furtherconfigured to perform a training pass to fill FIR filter coefficientswith the samples before the FIR adaptation begins.
 6. The radar sensingsystem of claim 5, wherein the training pass is run backwards so thatthe FIR filter coefficients at the end of the training corresponds to achirp frequency at the beginning of the receive window.
 7. A method formitigating frequency modulated interference in a radar sensing system,the method comprising: providing a plurality of transmitters configuredfor installation and use on a vehicle; providing a plurality ofreceivers configured for installation and use on the vehicle;transmitting, with the transmitters, phase-modulated radio signals;receiving, with the receivers, radio signals, wherein the received radiosignals include (i) the transmitted radio signals transmitted by thetransmitters and reflected from objects in the environment, and (ii)interfering frequency-modulated radio signals transmitted by anotherradar sensing system; controlling the transmitters and the receiverssuch that the transmitters and the receivers operate in alternatingtransmit and receive windows, wherein the transmit windows are separatedin time from the receive windows; wherein each receiver of the pluralityof receivers comprises a linear frequency modulation canceler; andmitigating, with a linear frequency modulation canceler, the interferingradio signals received by an associated receiver, such that theassociated receiver is able to estimate a location of an object; andwherein the linear frequency modulation canceler comprises a finiteimpulse response (FIR) filter and is configured as a 1-step linearpredictor with least mean squares adaptation, wherein the prediction issubtracted from an FIR input signal to generate an error signal thatdrives the adaptation, and wherein the linear frequency modulationcanceler generates the canceler output.
 8. The method of claim 7 furthercomprising controlling the adaptation on a first receiver of theplurality of receivers, and copying FIR coefficients from the firstreceiver to the other receivers of the plurality of receivers, whereinthe adaptation is a FIR adaptation.
 9. The method of claim 7, whereinthe FIR filter comprises a maximum of 15 taps.
 10. The method of claim 7further comprising delaying the adaptation such that transients at thestart of each receive window are avoided.
 11. The method of claim 10further comprising delaying the adaptation until a delay line of the FIRfilter has filled with valid signal samples, and performing a trainingpass to fill FIR filter coefficients with the samples before theadaptation begins.
 12. The method of claim 11, wherein the training passis run backwards so that the FIR filter coefficients at the end of thetraining corresponds to a chirp frequency at the beginning of thereceive window.
 13. A radar sensing system comprising: a transmitterconfigured to transmit phase-modulated continuous wave radio signals; areceiver configured to receive radio signals that include (i) thetransmitted radio signals transmitted by the transmitter and reflectedfrom objects in the environment, and (ii) interferingfrequency-modulated radio signals transmitted by another radar sensingsystem; a controller configured to control the transmitter and thereceiver such that the transmitter and the receiver operate inalternating transmit and receive windows, wherein the transmit windowand the receive window are separated in time; wherein the receivercomprises a frequency modulation canceller configured to cancel asubstantial portion of the interfering radio signals received by thereceiver, such that the receiver is able to estimate a location of anobject; and wherein the frequency modulation canceler is a linearfrequency modulation canceler, and wherein the linear frequencymodulation canceler comprises a finite impulse response (FIR) filter andis configured as a 1-step linear predictor with least mean squaresadaptation, wherein the prediction is subtracted from an FIR inputsignal to generate an error signal that drives the adaptation, andwherein the linear frequency modulation canceler generates the canceleroutput.
 14. The radar sensing system of claim 13, wherein theinterfering radio signals are modulated with a linear frequencymodulation.
 15. The radar sensing system of claim 13, wherein thecontroller delays the adaptation such that transients at the start ofeach receive window are avoided.
 16. The radar sensing system of claim13, wherein the controller is configured to delay the FIR adaptationuntil a delay line of the FIR filter has filled with valid signalsamples, further configured to perform a training pass to fill FIRfilter coefficients with the samples before the FIR adaptation begins,and wherein the training pass is run backwards so that the FIR filtercoefficients at the end of the training corresponds to a chirp frequencyat the beginning of the receive window.